The present invention relates to a switching power supply circuit provided as a power supply for various electronic apparatus.
Various switching power supply circuits formed with various resonance type converters, for example, have been proposed. Resonance type converters readily obtain high power conversion efficiency and achieve low noise by forming a sinusoidal waveform in switching operation. The resonance type converters have another advantage of being able to be formed by a relatively small number of parts.
FIG. 14 is a circuit diagram showing an example of a prior art switching power supply circuit. As a fundamental configuration of the power supply circuit shown in FIG. 14, a voltage resonance type converter is provided as a switching converter on the primary side.
In the power supply circuit shown in FIG. 14, a bridge rectifier circuit Di and a smoothing capacitor Ci generate a rectified and smoothed voltage Ei from a commercial alternating-current power.
The voltage resonance type converter for receiving and interrupting the rectified and smoothed voltage Ei employs a single-ended system using one transistor. A self-excited configuration is employed as a driving system. In this case, a bipolar transistor such as a high withstand voltage BJT (Bipolar Junction Transistor) is selected as a switching device Q1 for forming the voltage resonance type converter. A primary-side parallel resonant capacitor Cr is connected in parallel with a collector and an emitter of the switching device Q1. A clamp diode DD is connected between a base and the emitter of the switching device Q1. The parallel resonant capacitor Cr forms a primary-side parallel resonant circuit in conjunction with leakage inductance L1 obtained at a primary winding N1 of an isolation converter transformer PIT, whereby operation of the voltage resonance type converter is obtained.
The base of the switching device Q1 is connected with a self-oscillation driving circuit comprising a driving winding NB, a resonant capacitor CB, and a base current limiting resistance RB. The switching device Q1 is supplied with a base current based on an oscillating signal generated in the self-oscillation driving circuit, and is thereby driven for switching operation. Incidentally, at the time of a start, the switching device Q1 is started by a starting current flowing from a line of the rectified and smoothed voltage Ei to the base of the switching device Q1 via a starting resistance Rs.
FIG. 15A and FIG. 15B show a structure of an orthogonal type control transformer PRT. FIG. 15A is an external perspective view of assistance in explaining a general structure of the orthogonal type control transformer PRT. FIG. 15B is a sectional perspective view of assistance in explaining winding directions of windings wound in the orthogonal type control transformer PRT.
The orthogonal type control transformer PRT is formed by winding a control winding Nc in a winding direction orthogonal to a winding direction of the driving winding NB and a resonance current detecting winding ND.
The orthogonal type control transformer PRT has a gap length G of 10 xcexcm at junctions of magnetic legs 21a to 21d and magnetic legs 22a to 22d, respectively.
The control winding Nc of the orthogonal type control transformer PRT is formed by a 60 xcexcm xcfx86 polyurethane-covered copper wire wound by 1000 T (turns), for example; the detecting winding ND is formed by a 0.3 mm xcfx86 polyurethane-covered copper wire wound by 1 T; and the driving winding NB is formed by a 0.3 mm xcfx86 polyurethane-covered copper wire wound by 3 T.
The isolation converter transformer PIT transmits a switching output of the switching converter obtained on the primary side to the secondary side.
As shown in FIG. 16, for example, the isolation converter transformer PIT has an Exe2x80x94E-shaped core formed of E-shaped ferrite cores CR1 and CR2. As shown in FIG. 16, the primary winding N1 and a secondary winding N2 each formed by a litz wire are wound in respective divided regions using a dividing bobbin B.
A gap G is formed in a central magnetic leg of the Exe2x80x94E-shaped core, as shown in FIG. 16. Gap length of the gap G determines leakage inductance in the isolation converter transformer PIT. Also, loose coupling at a required coupling coefficient is obtained by the gap length of the gap G. The coupling coefficient in this case is 0.85, for example, to obtain a state of loose coupling, and accordingly saturation is not readily reached. The gap G can be formed by making the central magnetic leg of the E-shaped cores CR1 and CR2 shorter than two outer magnetic legs of the E-shaped cores CR1 and CR2. The gap length in this case is about 1 mm.
As shown in FIG. 14, the primary winding N1 of the isolation converter transformer PIT has one end connected to the line of the direct-current input voltage (rectified and smoothed voltage Ei) via the current detecting winding ND, and another end connected to the collector of the switching device Q1. The switching device Q1 performs switching operation on the direct-current input voltage. With the above-described form of connection, the switching output of the switching device Q1 is supplied to the primary winding N1 and the current detecting winding ND, and thus an alternating voltage having a cycle corresponding to switching frequency occurs.
An alternating voltage induced by the primary winding N1 of the isolation converter transformer PIT occurs in the secondary winding N2. In this case, a secondary-side parallel resonant capacitor C2 is connected in parallel with the secondary winding N2. Thereby, leakage inductance L2 of the secondary winding N2 and capacitance of the secondary-side parallel resonant capacitor C2 form a parallel resonant circuit. The parallel resonant circuit converts the alternating voltage induced in the secondary winding N2 to a resonance waveform. That is, a voltage resonance operation is obtained on the secondary side.
On the secondary side of the isolation converter transformer PIT in this case, an anode of a rectifier diode D01 is connected to the secondary winding N2 and a cathode of the rectifier diode D01 is connected to a smoothing capacitor C01, whereby a half-wave rectifier circuit is formed. The half-wave rectifier circuit provides a secondary-side direct-current output voltage E01 across the smoothing capacitor C01.
Further, in this case, the secondary winding N2 is provided with a tap. As shown in FIG. 14, a half-wave rectifier circuit comprising a rectifier diode D02 and a smoothing capacitor C02 is formed for the tap output. The half-wave rectifier circuit provides a secondary-side direct-current output voltage E02 lower than the secondary-side direct-current output voltage E01.
The secondary-side direct-current output voltages E01 and E02 are each supplied to a required load circuit. The secondary-side direct-current output voltage E01 is also outputted from a branch point as a detection voltage for a control circuit 1.
The control circuit 1 functions as an error amplifier receiving the direct-current output voltage E01 as a detection input. Specifically, a voltage obtained by dividing the direct-current output voltage E01 by resistances R3 and R4 is inputted as a control voltage to a control terminal of a shunt regulator Q3. Hence the shunt regulator Q3 allows a current having a level corresponding to the direct-current output voltage E01 to flow as a control current Ic to the control winding Nc. That is, the level of the control current flowing through the control winding Nc is variably controlled.
Since the level of the control current flowing through the control winding Nc is changed, the orthogonal type control transformer PRT effects control so as to change inductance LB of the driving winding NB. Thereby resonance frequency of a resonant circuit comprising the driving winding NB and the resonant capacitor CB in the self-oscillation driving circuit is changed, and therefore the switching frequency of the switching device Q1 is variably controlled. Since the switching frequency of the switching device Q1 is thus changed, the secondary-side direct-current output voltage is controlled and stabilized to be constant. Incidentally, the inductance LB of the driving winding NB changes from 8 xcexcH to 2.5 xcexcH for the control current Ic=10 mA to 60 mA.
The orthogonal type control transformer PRT has a very small gap G of about 10 xcexcm as described above in order to reduce the amount of control current to be passed through the control winding. Thus, at the time of manufacturing, errors in accuracy of gap thickness of the gap G occur, causing variations in the inductance value of the driving winding NB wound in the orthogonal type control transformer PRT.
Variations in permeability of the ferrite cores, displacement between the magnetic legs at the time of joining, and the like also result in variations in the inductance value of the driving winding NB.
As a result of these, the value of the inductance LB varies about xc2x110%.
In addition, winging the winding NC in a direction orthogonal to that of the detecting winding ND and the driving winding NB in the orthogonal type control transformer PRT greatly complicates a winding process in manufacturing. Further, joining the magnetic legs of the cores 21 and 22 together without displacement makes an assembly process difficult. Thus, the orthogonal type control transformer PRT is manufactured with a high degree of difficulty, and its cost is difficult to reduce.
In view of the above problems, a switching power supply circuit according to the present invention is comprised as follows.
The switching power supply circuit includes: switching means including a first switching device for performing switching operation on a direct-current input voltage; an isolation converter transformer connected in series with the first switching device, for transmitting a switching output obtained in the primary winding by the switching operation to a secondary winding; and a first resonant circuit formed by the primary winding of the isolation converter transformer and a first capacitor, for producing voltage resonance of the switching output.
The switching power supply circuit further includes a drive transformer having a detecting winding connected in series with the primary winding or the secondary winding of the isolation converter transformer, a driving winding excited by a switching output obtained in the detecting winding, and a control winding for controlling inductance of the driving winding by a change in current level, at least the driving winding and the control winding being wound on an identical core.
The switching power supply circuit further includes switching driving means having a second resonant circuit formed by the driving winding and a second capacitor, for switching driving of the first switching device on the basis of an output of the second resonant circuit.
The switching power supply circuit further includes direct-current output voltage generating means for rectifying the switching output transmitted to the secondary winding and thereby providing a direct-current output voltage.
The switching power supply circuit further includes constant-voltage control means having a series connection circuit formed by connecting a second switching device in series with the control winding, for variably controlling switching frequency of the first switching device by variably controlling the current level in the series connection circuit according to a level of the direct-current output voltage, and thereby effecting constant-voltage control on the direct-current output voltage.
The power supply circuit thus comprised has the drive transformer. The power supply circuit thus employs a fundamental configuration of a complex resonance type converter driving the switching device by self-excitation. For constant-voltage control, the power supply circuit has the series connection circuit including the second switching device. A current flowing through the switching driving means for performing switching driving by self-excitation branches to the series connection circuit via the control winding of the drive transformer. By varying the current level in the series connection circuit, the amount of current flowing through the switching driving means is changed, whereby the switching frequency of the switching device is variably controlled.
With such a constant-voltage control configuration, variations in the inductance value when the orthogonal type control transformer that has been in use for variably controlling the switching frequency is used in the case of self-excitation, for example, can be prevented.
Further, in place of such a drive transformer, a driving winding is provided to an isolation converter transformer, a second resonant circuit formed by the driving winding, an inductor, and a capacitor is formed, and switching driving means for switching driving of the first switching device on the basis of an output of the second resonant circuit is provided. A second switching device is connected in parallel with the switching driving means. A current level in the second switching device is variably controlled according to a level of a direct-current output voltage. It is thereby possible to variably control switching frequency of the first switching device, and thus effect constant-voltage control on the direct-current output voltage.